Timing recovery using the pilot signal in high definition TV

ABSTRACT

Improved carrier recovery and symbol timing systems and methods suitable for use in connection with a dual-mode QAM/VSB receiver system is disclosed. Carrier and symbol timing acquisition and tracking loops are phase/frequency locked to an inserted pilot signal provided in an input VSB spectrum at a given frequency. An input spectrum is centered about baseband and the pilot is extracted by an equivalent filter which functions as a bandpass filter having pass bands centered about the pilot frequency. Since the pilot signal&#39;s frequency is given, its position in the frequency domain for any sampling frequency, is deterministic. The receiver&#39;s sampling frequency is provided such that the relationship is expressed as f C =f S   4 . When tracked by a phase-lock loop, the pilot signal will appear at the correct location in the spectrum if the sampling frequency f S  is correct, and will be shifted in one direction or the other if the sampling frequency f S  is too high or too low.

PRIORITY CLAIM

The present application claims the benefit of the priority date of U.S.Provisional Applications Serial Nos. 60/106,921, filed Nov. 3, 1998,60/106,922, filed Nov. 3, 1998, 60/106,923, filed Nov. 3, 1998,60/106,938, filed Nov. 3, 1998, 60/107,103, filed Nov. 4, 1998 and60/107,037, filed Nov. 3, 1998, the entire disclosures of which areexpressly incorporated herein by reference and also is a continuation ofSer. No. 09/433,734 filed Nov. 3, 1999.

FIELD OF THE INVENTION

The present invention relates to systems for, and methods of, recoveringdigitally modulated television signals and, more particularly, to a dualmode QAM/VSB receiver system for recovering quadrature amplitudemodulated or vestigial sideband modulated signals.

BACKGROUND OF THE INVENTION

Modern digital telecommunication systems are operating atever-increasing data rates to accommodate society's growing demands forinformation exchange. However, increasing the data rates, while at thesame time accommodating the fixed bandwidths allocated by the FederalCommunications Commission (FCC), requires increasingly sophisticatedsignal processing techniques. Since low cost, small size and low powerconsumption are portent in the hardware implementations of suchcommunication systems, custom integrated circuit solutions are importantto achieving these goals.

Next generation digital television systems, such as cable transportedtelevision (CATV) and high-definition television (HDTV) rely ontelecommunication transceivers to deliver data at rates in excess of 30megabits per second (30 Mb/s). The ATSC A/53 Digital TelevisionStandard, was developed by the “Digital HDTV Alliance” of U.S.television vendors, and has been accepted as the standard forterrestrial transmission of SDTV and HDTV signals in the United States.The ATSC A/53 standard is based on an 8-level vestigal sideband (8-VSB)modulation format with a nominal payload data rate of 19.4 Mbps in a 6MHz channel. A high data rate mode, for use in a cable televisionenvironment, is also specified by the standard. This particular mode,defined in Annex D to the ITU-T J.83 specification, utilizes a 16-VSBmodulation format to provide a data rate of 38.8 Mbps in a 6 MHzchannel.

Transmission modes defined in ITU-T J.83 Annex A/C are used primarilyoutside the United States for digital cable television transmission. Thetransmission modes supported by this specification have been adopted inEurope as the Digital Video Broadcast for Cable (DVB-C) standard, andfurther adopted by the Digital Audio-Video Council (DAVIC) withextensions to support 256-QAM modulation formats.

Beyond these divergent requirements, the ITU-T J.83 Annex B standardsdefine the dominant methodology for digital television delivery overCATV networks in the United States. It has been adopted as the physicallayer standard by various organizations including the SCTE DVS-031,MCNS-DOCSIS and the IEEE 802.14 committee.

Given the implementation of multiple modulation techniques in thevarious adopted standards, there exists a need for a television receiversystem capable of receiving and demodulating television signalinformation content that has been modulated and transmitted inaccordance with a variety of modulation formats. In particular, such asystem should be able to accommodate receipt and demodulation of atleast 8 and 16-VSB modulated signals in order to support US HDTVapplications, as well as 64 and 256-QAM modulated signals, for Europeanand potential US CATV implementations.

SUMMARY OF THE INVENTION

The present invention is directed to digital data communication systemsand methods for operating such systems in order to synchronize areceiver's timebase to a remote transmitter's. Carrier frequency andsymbol timing information is recovered from a pilot (unsuppressedcarrier) signal that is inserted into a VSB spectrum. Unlikeconventional timing recovery systems, which recover timing informationfrom the segment sync signal, that is provided at the end of every lineof 828 symbols, and is specifically designed to facilitate timingrecovery.

In a first aspect of the invention, a digital communication systemincludes an analog front end which receives an input spectrum in anintermediate frequency. The input spectrum includes an inserted pilotsignal, representing a pre-determined frequency component. First andsecond nested tracking loops are provided, with the first loop acquiringcarrier frequency lock in operative response to the predeterminedfrequency component. The second loop provides a signal adapted toposition the input spectrum at a predetermined location relative tobaseband in operative response to the predetermined frequency component.A third tracking loop is coupled so as to define a symbol timingparameter in operative response to the same predetermined frequencycomponent. The digital communication system includes an equivalentfilter which operates on the received spectrum in order to define a pairof symmetric signals, each of the symmetric signals centered at thecharacteristic frequency of the predetermined frequency component whenthe received spectrum is at baseband.

In an additional aspect of the invention, the equivalent filter isconstructed of a first, high pass filter, having a lower cut-offfrequency which is related to the system's sampling frequency. Theequivalent filter further includes a second, low pass, filter which hasan uppercut-off frequency bearing the same relationship to the samplingfrequency as the high pass filter. The first and second filterstherefore define an equivalent band pass filter having symmetricpassband regions centered about a frequency bearing the samerelationship to the sampling frequency. When the received spectrumexhibits a raised cosine response characteristic, the passband regionsincorporate the transition regions of the high and low pass filteredspectra.

The pilot signal is provided at a characteristic predetermined frequencyf_(C). In one particular embodiment of the invention, the samplingfrequency f_(S) is chosen such that the pilot frequency is equal tof_(S)/4. The high pass filter accordingly has a lower cut-off of f_(S)/4and a passband center of about f_(S)/2.The low pass filter has an uppercut-off of about f_(s)/4, the equivalent filter passband is therebycentered at a frequency of f_(S)/4. Since one-fourth the samplingfrequency, i.e., f_(S)/4, is designed to be equal to the pilot frequencyf_(C), the equivalent filter passband regions are centered at f_(C) whenf_(C)=f_(S)/4.

In a further aspect of the invention, an equivalent filter passbandsignal is provided to a phase/frequency detector which is constructed soas to determine whether the pilot signal is centered in the passbandregion. An oscillator circuit develops a timing reference signal havinga frequency related to the sampling frequency, the oscillator circuitincreasing or decreasing the timing reference signal frequency inoperative response to the position of the pilot signal with respect tothe passband region center.

A system according to the invention might thus be characterized asincluding a filter circuit for isolating an inserted pilot signal; adetector circuit coupled to receive the isolated pilot signal andcompare its frequency value to a predetermined frequency; a frequencyreference generation circuit for increasing or decreasing a referencefrequency based upon the comparison result and a symbol timing circuit,defining consecutive symbol occurrence intervals, in operative responseto the reference frequency. The symbol timing circuit operates at asample frequency having an integer relationship to the pilot signal. Thepilot signal will appear at a correct location in the spectrum if thesampling frequency is correct. The pilot signal will be shifted awayfrom its expected frequency location, in a first direction, if thesampling frequency is too high, and will be shifted away from theexpected frequency location, in the other direction, if the samplingfrequency is too low.

The equivalent filter passband signals, containing an augmented pilot,are provided as inputs not only to a symbol timing loop, but also tofirst and second carrier recovery loops. Carrier recovery and symboltiming is therefore performed in operative response to the sameaugmented pilot input signal.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other features, aspects and advantages of the presentinvention will be more fully understood when considered with respect tothe following detailed description, appended claims and accompanyingdrawings wherein:

FIG. 1 is a simplified, semi-schematic block diagram of a dual modeQAM/VSB receiver architecture in accordance with the invention;

FIG. 2 is a graphical representation of QAM, VSB and offset-QAM spectrasubtended by their respective eye diagrams illustrating both the I and Qrails;

FIG. 3 illustrates a typical 6 MHz spectra represented as a raisedcosine waveform illustrating transition regions and the location of apilot signal;

FIG. 4 is a simplified, semi-schematic block diagram of the architectureof carrier recovery and baud loops of a dual mode QAM/VSB receiver inaccordance with the invention;

FIG. 5 is a simplified, semi-schematic block diagram of a square rootNyquist low pass filter in combination with a Nyquist high passprefilter expressed as an equivalent bandpass filter;

FIG. 6 is a graphical representation of the affects of the low pass,high pass and equivalent bandpass filters of FIG. 5 on an inputspectrum, where the filter's cutoff frequencies have an integerrelationship to the sampling frequency;

FIG. 7 is a simplified, semi-schematic block diagram of a baud loop asmight be implemented in a dual mode QAM/VSB receiver architecture inaccordance with the invention;

FIG. 8 is a simplified, semi-schematic block diagram of a phase detectoras might be implemented in the baud loop of FIG. 7;

FIG. 9 is a simplified, semi-schematic block diagram of a dual modeQAM/VSB receiver architecture, including decision directed carrier phasetracking circuitry in accordance with the invention;

FIG. 10 is a simplified, semi-schematic block diagram of a QAM phasedetector suitable for implementation in the dual mode QAM/VSB receiverarchitecture of FIG. 9;

FIG. 11 is a simplified, semi-schematic block diagram of a VSB phasedetector, where the Hilbert transform of an input signal is provideddirectly;

FIG. 12 is a simplified, semi-schematic block diagram of a VSB phasedetector where the Hilbert transform of an input signal is providedwithin the phase detector;

FIG. 13 is a simplified, semi-schematic block diagram of a single bitderotater provided at the equalizer input of the dual mode QAM/VSBsystem of FIG. 9;

FIG. 14 is a simplified, semi-schematic block diagram of a decisionfeedback equalizer;

FIG. 15 is a simplified block diagram of an exemplary 8-tap decisionfeedback filter;

FIG. 16 is a simplified semi-schematic block diagram of a complexdecision feedback or complex decision feedforward filter;

FIG. 17 is a graphical representation of a 256 QAM constellation;

FIG. 18 is a simplified, semi-schematic block diagram of a decisionfeedback equalizer including computational offset correction circuitryin accordance with the invention configured for QAM modulated signals;

FIG. 19 is a simplified, semi-schematic block diagram of a decisionfeedback equalizer including a pilot tone generation circuit;

FIG. 20 is a simplified, semi-schematic block diagram of a decisionfeedback equalizer in accordance with the invention including offsetcorrection circuitry for VSB modulated signals;

FIG. 21 is a simplified, semi-schematic block diagram of a trellisencoder including a symbol mapper suitable for 8 VSB transmission;

FIG. 22 is a simplified, semi-schematic block diagram of a decisionfeedback equalizer circuit, including carrier and timing loops and asymbol-by-symbol slicer;

FIG. 23 is a simplified, semi-schematic block diagram of a decisionfeedback equalizer circuit, including carrier and timing loops and a TCMdecoder circuit in accordance with the invention;

FIG. 24 is a simplified, semi-schematic block diagram of a decisionfeedback equalizer circuit depicting the construction and arrangement ofa TCM decoder circuit in accordance with the invention;

FIG. 25 is a simplified, semi-schematic block diagram of a 4-statetraceback path memory circuit suitable for practice of the presentinvention.

DETAILED DESCRIPTION OF THE INVENTION

One particular aspect of the present invention might be implemented in adual mode QAM/VSB receiver system such as illustrated in simplified,semi-schematic block diagram form in FIG. 1. The receiver systemillustrated in FIG. 1 can be described as a digital receiver which iscompatible with both North American digital cable television and digitalterrestrial broadcast television standards. The receiver system of FIG.1 is capable of receiving all standard-definition and high-definitiondigital television formats (SDTV/HDTV).

In accordance with principles of the invention, the receiver systemdepicted in FIG. 1 accepts an analog signal centered at standardtelevision IF frequencies, amplifies and digitizes the input analogsignal with an integrated programmable gain amplifier and 10-bit A/Dconverter. Digitized signals are demodulated and filtered with acombined 64/256-QAM and 8/16-VSB demodulator and are adaptively filteredto remove multipath propagation effects and NTSC co-channelinterference. The resulting digital data is error corrected withintegrated trellis and Reid-Solomon decoders which support both the ATSCA/53 and ITU T J.83 Annex A/B/C coding formats. The final receive datastream is delivered in either parallel or serial MPEG-2 transport formatfor displaying on a television screen. It should be noted that thereceiver system of FIG. 1 is suitable for digital CATV/HDTV set-top boxapplications as well as digital CATV/HDTV televisions.

In the exemplary embodiment of the receiver of FIG. 1, all clock,carrier, gain acquisition and tracking loops are integrated with thedemodulation and decoding functionality on a single integrated circuitchip, as are the necessary phase-locked-loops, referenced to a singleexternal crystal.

The analog front end of the dual mode QAM/VSB receiver of FIG. 1,indicated generally at 10, suitably includes a programmable gainamplifier (PGA) 12 and a 10-bit analog-to-digital (A/D) converter 14.The PGA 12 is controlled by an on-chip gain recovery loop, operating inconventional fashion, to implement an automatic gain control (AGC)function. The A/D converter 14 is clocked by an on-chip voltagecontrolled oscillator (VCO) which is locked to an off-chip crystalresonator functioning as a stable timing reference. This stablereference allows an input intermediate frequency (IF) signal to besubsampled in order to produce a digital data stream centered on asubstantially lower IF center frequency.

Digressing momentarily, it should be noted that the dual mode QAM/VSBreceiver of FIG. 1 contemplates supporting two modes of IF inputsignals, direct sampling of a QAM spectrum centered on a low IF, orsubsampling of a QAM spectrum centered on a standard tuner IF frequencyof 44 Megahertz (MHz). In low IF mode, the output of a conventionaltuner is first passed through a 6 MHz SAW filter centered on the tunerIF frequency to limit out-of-band signal energy. The differential SAWoutput is then AC coupled to a conventional downconversion circuit whichcenters the QAM spectrum on a low IF such as 6 MHz, and amplifies itunder control of the AGC 12 to provide a nominal 1.0 volt peak-to-peaksignal.

Returning now to the exemplary embodiment of FIG. 1, the dual modeQAM/VSB receiver according to the invention further includes an IF DCoffset cancellation circuit 16 which compensates for any DC shiftintroduced by the A/D circuit 14. A complex mixer 18 subsequentlyconverts IF sample data into baseband data and is controlled by a directdigital frequency synthesizer (DDFS) driven by the carrier frequencyrecovery loop in a manner to be described in greater detail below.

The QAM/VSB receiver's demodulator section suitably incorporates thecomplex digital mixer 18 and a multi-rate filter/interpolator (HB/VID)20 which in combination, converts an over sampled IF input signal to abaseband complex data stream which is correctly sampled in bothfrequency and phase, under control of a clock recovery loop, in a mannerto be described in greater detail below.

In-phase (I) and quadrature phase (Q) baseband signals are then filteredby square-root Nyquist filters 22 which can accommodate roll-off factorsof 11-18%. The outputs of the square-root Nyquist filters aresubsequently directed to an adaptive equalization block 24 and areparallel-processed by a Nyquist-type prefilter 26 to provide an inputsignal to an acquisition/tracking loop circuit 28 which includes carrierrecovery loop circuitry to Support carrier frequency recovery andspectrum centering as well as baud recovery loop circuitry, for symboltiming extraction, as will be described in greater detail below.

Prior to being directed to the Nyquist prefilter 26 and adaptiveequalization block 24, filtered signals are provided from thesquare-root Nyquist filter 22 to an NTSC co-channel interferencerejection filter 28, for removal of the luma, chroma, and audiosubcarrier signals from the frequency spectrum. When used in aterrestrial environment, there exists the possibility of co-channelinterference from terrestrial-type NTSC transmitters. The NTSCco-channel rejection filters 28 function as an adaptive digital filterwhich places precisely located notches in the frequency spectrum at thespecific locations of the NTSC luma, chroma, and audio subcarriers. AnNTSC co-channel rejection filter suitable for implementation inconnection with the dual mode QAM/VSB receiver system of FIG. 1, mightbe one such as described in co-pending patent application serial No.09/303,783, filed, May, 11, 1999 and entitled “NTSC REJECTION FILTER”,commonly owned by the Assignee of the present invention, the entiredisclosure of which is expressly incorporated herein by reference.

While the square-root Nyquist filters 28 ordinarily assure that there isa minimal inter-symbol interference (ISI) over a perfect channel, theNyquist filters are unable to remove ISI due to the imperfect channelcharacteristics. Accordingly, the dual mode QAM/VSB receiver accordingto the invention, provides an adaptive, multi-tap decision directedequalizer circuit 24, having 64 feedforward taps and 432 feedback taps,which is sufficient to remove ISI components generated by worst-casecoaxial cable and terrestrial broadcast channels with multi-path spreadsof up to 40 μsec at 10.76 Mbaud.

Blind convergence algorithms are utilized by the equalizer 24 along withan ability to implement a training sequence embedded in the incomingdata stream. In addition to adaptive equalization, the decision directedequalizer 24 also includes particular circuitry to perform carrierfrequency acquisition and phase tracking (in the case of QAM modulation)or carrier phase tracking (phase recovery in the case of VSB modulation)on equalized constellation points using a quadrature synthesizer andcomplex mixer under control of the carrier recovery loop, to track outresidual carrier offsets and instantaneous phase offsets such as arecaused by tuner microphonics, as will be described in greater detailbelow.

The dual mode QAM/VSB receiver exemplified in FIG. 1 further includes aforward error correction (FEC) and decoder block 32, which is compatiblewith all common CATV standards and the ATSC terrestrial broadcaststandard. Specifically, the Annex A/C decoder circuitry implements fourgeneral functions, frame synchronization, convolutional deinterleaving,reed-Solomon error correction and derandomization. Hard decisions areinput to the frame synchronizer which locks onto the inverted synch bytepattern, conventionally provided in television data frames. Aftersynchronization, data interleaving is removed by a convolutionaldeinterleaver utilizing a Ramsey type III approach. Data symbols arenext provided to a Reed-Solomon decoder, which is able to correct up to8 symbol errors per RS block, followed by data derandomization to undothe corresponding randomization operation of the transmitter'smodulator.

In the Annex B mode, the decoder typically performs five generalfunctions, and differs from the Annex A/C case primarily in its use oftrellis decoding. Soft decisions from the receiver's equalizer circuitare input to a trellis decoder which functions as a maximum likelihoodsequence estimator. Output sequences are directed to a framesynchronizer and derandomization block, similar to those describedabove, in connection with Annex A/C decoding. Data then is directed to aReed-Solomon decoder block which is capable of correcting 3 symbolerrors per RS block. A checksum decoder identifies blocks withuncorrectable errors and flags an output MPEG-2 data stream with aTransport Error Indicator (TEI) flag.

In practice, the majority of communication with the dual mode QAM/VSBreceiver 10 of FIG. 1, and the majority of the activity provided by thevarious functional blocks, takes place upon initiation of a channelchange. Upon detection of a channel change request, a receiver system'shost microprocessor determines whether the existing 6 MHz channelcontains the requested MPEG service or if another channel must beselected. In the latter case, the host microprocessor typically consultsits program table and might direct the receiver system to program achannel tuner to select the appropriate channel frequency. The hostmicroprocessor might then download, to the receiver 10, any channelspecific configuration that might be required, such as the configurationof the receiver and FEC 32 for reception of either a terrestrial (VSB)or a cable (QAM) channel.

Following configuration download the receiver 10 must acquire lock, i.e.synchronize its acquisition and tracking loop circuitry 30 to thefrequency and phase of a remote transmitter. Receiver lock is amulti-step process which generally involves allowing the variousacquisitions/tracking loops to acquire lock in a predetermined manner.For example, the AGC loops are generally allowed to acquire first, inorder to ensure that the signal level at the input to the A/D converter14 is set appropriately. AGC bandwidths are initially set wide open inorder to minimize acquisition time and subsequently reduced to provideadequate tracking and minimal noise.

Carrier frequency acquisition and symbol timing (baud timing) aretypically enabled after the AGC loops have acquired lock. Depending onthe particular mode of operation (QAM or VSB), these may be obtainedjointly or in sequence. In a manner to be described in greater detailbelow, each loop is allowed to acquire by widening the appropriatebandwidths, thus allowing the loops to pull-in the signal, and graduallyreducing the bandwidth as lock is obtained. Once baud timing and carrierfrequency is acquired, a carrier phase loop is enabled. While thecarrier frequency loop is typically able to obtain a course phase lock,its ability to track instantaneous phase noise is compromised. A carrierphase loop provides a superior ability to track out phase noise. Oncethe receiver system 10 has obtained lock, recovered data is delivered tothe FEC decoder 32. The FEC 32 first obtains node synchronization (ifthere is a trellis decoder in the selected coding scheme), following byframe synchronization. With frame synchronization achieved,derandomization and deinterleaving are performed along with Reid-Solomondecoding. MPEG-2 transport stream synchronization is then achieved anddata is delivered to the output for display.

The carrier frequency/phase recovery and tracking loops are all-digitalloops which simultaneously offer a wide acquisition range and a largephase noise tracking ability. In accordance with the present invention,the loops use both pilot tracking and decision directed techniques inorder to estimate the angle and direction for phase/frequencycompensation. The loops are filtered by integral-plus-proportionalfilters, in which the integrator and linear coefficients of the filterare programmable to provide means for setting loop bandwidths. The baudrecovery loop includes a timing error discriminant a loop filter, anddigital timing recovery block which controls a digital resampler. As wasthe case with the carrier loops, the baud loop's timing errordiscriminant outputs a new value each baud which is filtered by adigital integral-plus-proportional filter featuring programmablecoefficients.

In accordance with the present invention, the dual mode QAM/VSB receiversystem 10 of FIG. 1 is configurable to be operable with both NorthAmerican digital cable television (QAM) and digital terrestrialbroadcast television (VSB) standards as well as performing in a dualQAM/VSB mode, it should further be understood that VSB broadcasts mightbe one of two separate types, a first, terrestrial broadcast mode(referred to as 8 VSB) which supports a payload data rate of about 19.28Mbps in a 6 MHz channel, and a second, high data rate mode (referred toas 16 VSB) which supports a payload data rate of about 38.57 Mbps. Bothof these modes are well understood and described in the ATSC digitaltelevision standard, put forth by the advanced television systemscommittee. VSB transmission inherently requires only processing anin-phase (I) channel signal which is sampled at the symbol rate. Incontrast, QAM transmission requires that the receiver process bothin-phase (I) channel signals and quadrature-phase (Q) channel signalswhich are sampled at its symbol rate, typically one half that of acomparable VSM.

A comparison of the spectral distribution of QAM modulated signals andVSB modulated signals is illustrated in FIG. 2. Each of the spectra, forthe QAM and VSB cases, are subtended by an “eye” diagram illustratingthe signal content for both the I and Q rails. Although the VSB spectrummight be viewed as the sum of a real spectrum and its Hilbert transform,the VSB spectrum might further be considered as a frequency shiftedOffset-QAM (OQAM) spectrum. Accordingly, the dual mode QAM/VSB receiver10 of FIG. 1 is configured, when in VSB mode, to treat VSB modulatedsignals as either VSB or as OQAM, depending on the desires of the systemconfiguration engineers.

FIG. 2 includes an Offset-QAM spectrum subtended by its corresponding“eye” diagram, in which the distinctive feature of OQAM is evident. Inparticular, signals on the Q rail are delayed by one half of a symbol,thus offsetting the Q rail, in time, from information on the I rail. Ascan be seen in the VSB and OQAM spectra of FIG. 2, and as moreparticularly evident in the VSB channel occupancy diagram of FIG. 3, thespectrum occupying a nominal 6.0 MHz channel is generally flat, exceptfor symmetrical band edge regions where a nominal square root raisedcosine response results in 620 kHz transition regions 36 and 38.

A pilot signal 40, typically a 50 kHz pilot signal, is added to thespectrum by a pilot insertion circuit, implemented in accordance withthe standard, in all transmitters. The pilot signal 40 is typicallyprovided at a spectral position, 310 kHz from the lower band edge, thatwas reserved for the suppressed carrier signal in conventional NTSCtransmissions. This suppressed carrier signal provided a frequencyreference signal to which NTSC receivers could lock and which was usedfor carrier recovery. The pilot is also termed “pilot tone” and(misleadingly) “carrier”.

Carrier recovery is conventionally performed by an FPLL synchronousdetector, which integrally contains both the frequency loop and aphase-locked loop in one circuit. The frequency loop provides widefrequency pull-in range of approximately +/−100 kHz while thephase-locked loop might be implemented with a narrower bandwidth, i.e.,typically less than 2 kHz. Further, in the ATSC digital televisionstandard, the recommended approach to recover symbol timing informationis to utilize a data segment sync signal that makes up a VSB datasegment, and which is inserted between every segment of 828 symbols. Therepetitive data segment sync signals are detected from amongsynchronously detected random data by a narrow bandwidth filter. Fromthe data segment sync signals, a properly phased 10.76 MHz symbol clockis conventionally created.

In accordance with the present invention, the dual mode QAM/VSB receiver10 of FIG. 1 recovers timing information from the pilot (unsuppressedcarrier) signal that is included with the VSB signal, whereas the ATSCspecification intends that the pilot signal be used only for carrierrecovery.

Turning now to FIG. 4, there is depicted in simplified, semi-schematicblock diagram form, an exemplary embodiment of a unitary carrier andrecovery and symbol timing loop architecture, termed “unitary” in thatboth functions, frequency acquisition and tracking and symbol timing(also termed “baud recovery”) are operable in response to the pilot(unsuppressed carrier) signal. In the embodiment of FIG. 4, an input IFspectrum is digitized by an analog-to-digital converter (A/D) and theresulting digital complex signal is directed to a complex mixer 50 whereit is combined with a complex signal having a characteristic frequencyf_(C) equal to the carrier frequency. The resulting complex signal isprocessed by a highband filter and variable rate interpolator,represented as a single processing block in the embodiment of FIG. 4,and denoted HB/VID 52. In a manner to be described in greater detailbelow, symbol timing is performed by a baud loop coupled to providesymbol timing information to the variable rate interpolator (VID)portion of the HB/VID filter 52. Following interpolation, baseband IFsignals are processed by a square root Nyquist filter which has aprogrammable roll off α of from about 11 to about 18%. The square rootNyquist filter 54 is further designed to have a particular cutofffrequency that has a specific relationship to the VSB pilot frequencyf_(C), when the VSB spectrum centers at DC. In a manner to be describedin greater detail below, this particular cutoff frequency is chosen tohave this particular relationship in order that both carrier recoveryand symbol timing recovery might be based on a VSB pilot frequencyenhancement methodology.

An NTSC rejection filter 56 is provided in the signal path in order thatinterference components represented by the luma, chroma and audiosubcarriers, present in NTSC terrestrial broadcast system signals, areremoved from the digital data stream prior to the data being directed tothe receiver system's equalizer. The NTSC rejection filter 56 is an alldigital, programmable notch filter, exhibiting quite narrow notches atspecific, predetermined frequencies that correspond to the luma, chromaand audio subcarrier peaks. Although the NTSC rejection filter 56 iscontemplated as functioning to remove unwanted NTSC co-channelinterference components, the characteristics and design of the NTSCrejection filter 56 are such that it may be used to remove any form ofinterference component having a deterministic relationship to aparticular input spectrum.

Following the filter bank, the input baseband signal is directed to asecond mixer 58 where it's combined with a correction signal, developedin a manner to be described in greater detail below, which ensures thatthe spectrum is appropriately centered about zero.

It will thus be understood that there are two stages to carrieracquisition, a first stage, termed “an outside stage” (or outside loop)provides for mixing the received digitized spectrum down to baseband andwhich might properly be termed “a tracking loop”, and a secondcorrection stage, termed “an inside loop”, which functions more as anacquisition loop and which provides a correction factor to the spectrumto make sure the spectrum is properly centered. In addition, thecorrection factor is “leaked” from the inside loop to the outside loopin order that the inside loop might be constructed with a widebandwidth, typically in the 100 kHz range in order to provide for fastacquisition. Correction factors are leaked to the outside loop such thatthe outside loop might be constructed with a relatively narrow bandwidthin order to provide for more accurate tracking capability. Once thecarrier has been acquired.

A carrier phase detector 60 is coupled to receive an input signal from aNyquist prefilter 62 coupled in turn to receive complex signal from anode between the second mixer 58 and the receiver's equalizer 64. TheNyquist prefilter 62 is constructed as a high pass filter with a cutoffat the same particular characteristic frequency as the cutoff designatedfor the low pass root Nyquist filter 54. The root Nyquist filter 54 andNyquist prefilter 62 function in combination to define an equivalentfilter that acts to define the pilot enhanced timing recoverycharacteristics of the receiver in accordance with the presentinvention. Complex, pre-filtered signals are directed to the input ofthe carrier phase detector which produces a 6-bit frequency errordiscriminant for use in the loop. The SGN function of these 6-bits areextracted and applied, simultaneously, to an inside loop filter 66 andan outside loop filter 68. The inside loop filter 66 drives an insidetiming reference circuit, such as a direct digital frequency synthesizer(DDFS) which might also be implemented as a voltage controlledoscillator (VCO) or a numerically controlled oscillator (NCO). Likewise,the outside loop filter 68 drives an outside timing reference circuit 72which might also be suitably implemented as a DDFS, VCO, or an NCO. Aswas mentioned previously, the outside, or centering, loop functions todefine a complex signal that might be expressed as sin Ω_(Ct) and cosΩ_(Ct), where Ω_(C) represents the pilot (carrier) frequency. Since thepilot (carrier) frequency f_(C) is given, its position in the frequencydomain, with respect to any sampling frequency f_(S) is deterministic.Therefore, if a receiver system wishes to lock its timing frequency to aparticular F_(S) that has a fixed relationship with a known F_(C), as inthe case of the ATSC standard signals, it need only apply a phase lockloop that tracks the pilot. Axiomatically, the pilot signal will appearat the correct location in the spectrum if the sampling frequency F_(S)is correct. The pilot signal will be shifted to a lower frequency fromits expected frequency location if the sampling frequency f_(S) is toohigh. Conversely, in the case where the sampling frequency f_(S) is toolow, the pilot signal will appear to have been shifted to a higherfrequency location from its expected frequency location in the spectrum.

A particular case which makes implementation of pilot enhanced carrierrecovery simpler, occurs when the sampling frequency f_(S) is selectedto be four times that of the pilot frequency f_(C), when a VSB spectrumis centered at zero. Thus, when the spectrum is centered, the pilotsignal will be expected to occur at f_(C). In accordance with practiceof the present invention, the inside and outside loops will be lookingfor the pilot to occur at a frequency of f_(S)/4. This particularimplementation is illustrated in the semi-schematic block diagram ofFIG. 5 and its corresponding spectrum diagrams of FIG. 6.

As mentioned previously, the receiver system incorporates a frequencymodulated square root Nyquist low pass filter 54 in combination with ahigh pass Nyquist prefilter 62, which in combination might be viewed asa single equivalent filter 74. Both the root Nyquist 54 and Nyquistprefilter 62 are constructed with cutoff frequencies of f_(S)/4. Thus,and as indicated in the spectrum diagrams of FIG. 6, the high passNyquist prefilter 62 gives a resultant high and low band spectrum witheach centered about F_(S)/2. When the spectra of the root Nyquist filter54 and Nyquist prefilter 62 are superposed (summed as would be the casewith an equivalent filter 74) the resultant signal is a symmetricalwaveform, centered at F_(S)/4, each of which are centered about andsymmetric with respect to the pilot when the pilot frequency f_(C) isequal to f_(S)/4. Accordingly, since the pilot signal is designed to becentered within a spectrum's transition band, the combination of theroot Nyquist filter 54 and Nyquist prefilter 62 define an equivalentfilter 74 that provides an output signal symmetric about the pilot whenthe pilot (carrier) has been appropriately acquired. It will thus beunderstood that when the equivalent filter output is symmetric about thepilot f_(C), the resulting waveform can be represented as a puresinusoidal signal for which only zero crossings need to be evaluated bythe carrier phase detector (60 of FIG.4). If the sampling frequencyf_(S) is too high, not only will the pilot signal be observed to appearat a lower frequency than its expected frequency location, but also thesymmetry of the resultant waveform from the “equivalent” filter 74 willalso be disturbed due to the non-center placement of pilot. Likewise,when the sampling frequency f_(S) is too low, the pilot signal will beobserved to appear at a higher frequency than its expected frequencylocation, also perturbing the symmetry of the equivalent filter's outputin a direction opposite the previous case. The carrier phase detector(60 of FIG. 4) evaluates the position of the pilot with respect to thesampling frequency and provides appropriate correction signals to theinside loop filter (66 of FIG. 4) and the outside loop filter (68 ofFIG. 4).

During initialization, one is able to make certain assumptions about thepilot signal since its frequency f_(C) position with respect to thespectrum is deterministic. Accordingly, while the inside or acquisition,loop is acquiring the pilot, the outside, or centering, loop assumesthat no frequency offset has been introduced to the spectrum and runsthe DDFS (or VCO, or NCO) in a “flywheel” mode. Since the IF inputsignal is centered at 6 MHz, the outside timing reference (72 of FIG. 4)also runs at 6 MHz until such time as the inside loop is able to acquirethe carrier and “leak” any frequency offset information so obtained tothe outside loop filter (68 of FIG. 4) for developing appropriatecontrol signals for the outside loop's timing reference (72 of FIG. 4).

Returning now to FIG. 4, the Nyquist prefilter 62 further provides acomplex input signal to a baud loop (also termed “symbol timing loop”)which provides symbol timing information to the variable rateinterpolator 52. The baud loop suitably includes a baud phase detectorcircuit 76 coupled, in turn, to a baud loop filter 78 which controlsoperation of a baud timing generation circuit 80 such as a DDFS, VCO orNCO.

A further implementation of a baud loop is illustrated in the simplifiedsemi-schematic top level block diagram of FIG. 7. The implementation ofthe baud loop depicted in FIG. 7 is generally similar to that depictedin the acquisition and tracking loop diagram of FIG. 4, and suitablyincludes a baud phase detector coupled to receive complex signals I_prefand Q_pref from the Nyquist prefilter. The baud phase detector 76 mightbe implemented as a timing error discriminant which outputs a new valueeach baud, in turn, filtered by a digital integral-plus-proportional lowpass filter 78. The filtered signal is summed at a summing node 80 withan offset word, denoted “baud frequency control word” or SCW, and isused to control operation of a baud numerically controlled oscillator(NCO) 82. The loop is updated once per baud, but only if a sine changeoccurred on either the I or Q decision data since the previous baud. Thesumming node 80 and frequency control word are provided in order toaccommodate the baud loop to any known offsets that might, for example,have been acquired from past history of communication between thereceiver and a particular remote transmitter unit.

FIG. 8 is a semi-schematic block diagram of an exemplary implementationof the baud phase detector 76 of FIG. 7. In the exemplary embodiment ofa baud phase detector of FIG. 8, the input to the baud loop is thecomplex signal output by the Nyquist prefilter, I_pref and Q_pref. Thesignal from the I rail might either be offset by one symbol periodthrough a delay element 90 or alternatively, be provided directly to theremaining circuit elements of the baud phase detector through aselection MUX 92. In the case where the baud phase detector isprocessing a VSB signal, one symbol delay is added for signals on the Irail. In the case where the baud phase detector is processing QAMsignals, there is no need to add a symbol delay for either of the I orthe Q rail, since the I and Q components of a symbol are aligned withina symbol period. It should also be noted that when VSB signals are beingprocessed as OQAM, a one symbol delay is added to the signals on the Irail by passing the I signals through the delay element 90. Delayselection is made by the MUX 92 in response to a QAM/OQAM (VSB) signalprovided by an off-board control microprocessor.

After input, the sign of the symbols on the I and Q rails is determinedby a first sign logic circuit 94. The sign of the input symbols is mixedin mixer 104 with the output of a second sign logic circuit 102 whichdetermines the sign of signals appearing on the I Q rails after theyhave been directed through two sequential delay elements 96 and 98. Athird sign logic circuit 100 disposed between the two delay elements 96and 98 provides an output signal to a second mixer 106 where it iscombined with the output of the first mixer 104. The output of thesecond mixer 106, for both the I and Q signal paths, is summed by asumming circuit 108 and provided to the baud loop's low pass filter (78of FIG. 7). It will be understood that the baud loop is updated once perbaud, but only if a sign change occurred on either the I or Q decisiondata since the previous baud. The signs of the two previous symbols areevaluated by sign logic circuitry 100 and 102, while the sign of thepresent symbol is evaluated by sign logic circuitry 94. For example, iftwo sequential symbols exhibited particular phase relationships suchthat they might be characterized as rotated in a positive direction(i.e., having a positive sign), and a subsequent symbol exhibiting aphase rotation in the opposite direction (having a negative sign), thecombination of the signs of the first and third symbols would combine inmixer 104 to give a negative, and combined in mixer 106 with the sign ofthe second symbol to further result in a negative. This result wouldindicate a transition from the prior symbol's phase relationships and betherefore directed to the loop filter for control of the baud NCO. Thetiming error discriminant, as represented by the signals output by thesumming junction 108, is therefore represented by values of −1, 0 or +1,with 0 representing either of the case where I and Q are aligned in timeand in phase, or the case where I and Q are not aligned with oneanother, regardless of their phase relationship with nominal. In thislast case, unalike symbols are not necessarily indicative of baudtiming, but rather another form of error which it is not the function ofthe baud loop to compensate.

Returning momentarily to the simplified architectural illustration of anexemplary embodiment of the dual mode QAM/VSB transceiver of FIG. 1, itwill be understood that while the square root Nyquist filters 22 willassure that there is no intersymbol interference (ISI) over a perfectchannel, they are unable to remove ISI components due to imperfectchannel characteristics. Accordingly, After the filter block representedby the HB/VID 20, root Nyquist 22 and NTSC 28 filters, the dual modeQAM/VSB receiver provides a decision directed equalizer, incorporatingboth a feedforward equalizer and decision feedback equalizer forremoving such ISI components. In the exemplary embodiment, the adaptedequalizer 24 might be constructed as a 496-tap decision directedequalizer with 64-real/16-complex feedforward (FFE) taps and432-real/108-complex feedback (DFE) taps, which is sufficient to removeISI generated by worst-case coaxial cable and terrestrial broadcastchannels. In addition to adaptive equalization, the adaptive equalizer24 also includes circuitry for performing phase recovery on equalizedconstellation points, by using a quadrature synthesizer and complexmixer under the control of a carrier recovery loop, in order to trackout residual carrier offsets and instantaneous phase offsets such as arecaused by tuner microphonics. Further, the adaptive equalizer 24 isimplemented such that the same hardware is configurable for either QAMor VSB applications, with a complex implementation being used for QAMand a real implementation being used for VSB.

In the case of QAM modulated signals, both carrier frequency and phaserecovery is performed by circuitry contained within the adaptiveequalizer block 24. In the case of VSB modulated signals, the equalizersection further includes circuitry for performing carrier phaserecovery. In particular, since the adaptive equalizer incorporatesdecision directed circuitry, it is quite amenable to decision directedrecovery techniques. For QAM, I and Q are coincident in time, so if Iand Q are mixed, both the decision vector and phase offset are directlyobtainable.

For VSB (or OQAM), I and Q are offset from one another by one symbolperiod. Accordingly, some phase rotation metric must be defined before Iand Q are directed to the equalizer. As was described above, inconnection with FIG. 8, the baud loop artificially puts I and Q inphase, by action of the initial delay stage (90 of FIG. 8) disposed onthe I rail.

Turning now to FIG. 9, there is depicted a simplified, semi-schematicblock level diagram of the exemplary dual mode QAM/VSB receiver,including details of the construction and arrangement of adaptiveequalizer 24 having decision directed VSB phase tracking and decisiondirected QAM frequency acquisition and phase tracking loops inaccordance with the present invention. As illustrated in the embodimentof FIG. 9, the adaptive equalizer includes a feedforward (FFE) block 110configured to receive symbol aligned complex signals centered inbaseband. The FFE 110 is suitably constructed as either a 64-tap realFFE, for VSB applications, or a 16-tap complex FFE when used inconnection with QAM modulated signals. Carrier phase alignment and/orcarrier frequency/phase alignment is performed in a mixer 112 whichreceives signals from the FFE 110 and combines them with a timingreference signal developed by a timing reference circuit 114 such as anumerically controlled oscillator (NCO) a voltage controlled oscillator(VCO) or a direct digital frequency synthesizer (DDDFS). Timed signalsare then provided to a slicer 116 operating in conjunction with adecision feedback (DFE) block 118 which, in combination, provide harddecision information on constellation states as well as errorinformation relating to differences between actual signal trajectoryrelative to an ideal signal trajectory.

Error signals developed in the equalizer are directed through either aQAM phase detector 120 or a VSB phase detector 122 depending on how theincoming signal has been modulated. Choosing between the QAM phasedetector 120 and VSB phase detector 122 is a function of a multiplexcircuit 124 operating in response to a QAM/VSB control signal providedby an off-chip microprocessor. A second multiplex circuit 126 couplesthe output of the QAM phase detector 120 and VSB phase detector 122 to alow pass filter 128 which, in turn, develops control signals for thetiming reference circuit 114. Thus, the dual mode QAM/VSB receiver canbe characterized as having four separate and distinct timing loops,operative in various combinations depending on whether the incomingsignal is VSB or QAM.

In particular, the multiple loop timing system includes a first loop,also termed an inside loop, suitably including the Nyquist prefilter 62,carrier phase detector 60, an inside loop filter 66 and an inside timinggeneration circuit 70 such as an NCO, VCO or DDFS. The inside loopfunctions as a wide bandwidth acquisition loop for frequency recovery onthe carrier signal in a manner described above. The multiple loop systemfurther includes a second loop, also termed the outside loop, whichshares the carrier phase detector 60 with the inside loop and whichfunctions as a narrow bandwidth centering loop, also for frequencyrecovery on the pilot signal. The third loop, of the multiple loopsystem, is the baud loop which functions to define symbol timing. As wasdescribed above, the first loop, the inside or acquisition loop, isoperative only when the received signal is a VSB signal. The fourthloop, of the multi loop system including QAM and VSB phase detectors 120and 122 in combination with Lopez filter 128 and exemplary NCO 114,functions as the frequency recovery acquisition loop in the QAM case, aswell as the phase tracking loop for both VSB and QAM cases. As wasdescribed in connection with the inside and outside loops, above, thephase corrections developed by the QAM phase detector 120 are “leaked”to the outside loop's loop filter 68 on a 1-bit per period basis so asto provide a coarse correction to the outside loop in order that theoutside loop can be constructed with a narrow bandwidth in order tomaintain centering accuracy.

In the exemplary carrier recovery loop of FIG. 9, a particular form ofphase detection employs symbols with the same time stamps for each phaseerror term, thus allowing acquisition and tracking of carrier frequencyoffset in addition to tracking of carrier phase offset. Conventionally,VSB systems use every second consecutive symbol for phase detection,with the two symbols representing one symbol offset. Due to thisparticular time offset, the resultant carrier loop is insufficient toperform carrier frequency acquisition and tracking, as well as moresusceptible to self phase noise, introduced during phase detection whencompared to an equivalent QAM phase detection.

FIG. 10 is a simplified block diagram of a carrier recovery network suchas might be implemented in a dual mode QAM/VSB receiver. The carrierrecovery network includes a phase detector 130 configured to receivein-phase I and quadrature-phase Q input signals. The in-phase signalshould have been sampled twice a symbol at both the in-phase symbolsampling time and at the quadrature-phase sampling time. The in-phasesignal is then 1-to-2 de-multiplexed to generate two informationstreams, denoted I and X_(I), where the I represents in-phase symbolssampled at the in-phase sampling time and the X_(I) representsmid-symbol points sampled at the quadrature-phase symbol sampling time.

Similarly, the quadrature-phase symbol Q having been sampled twice asymbol is 1-to-2 de-multiplexed to generate two information streamsrepresenting the quadrature-phase symbols (Q) and its mid-point symbols(X_(Q)), respectively. Accordingly, when an in-phase signal isde-multiplexed in order to generate a symbol (I), the quadrature-phasesignal is de-multiplexed to generate its mid-symbol point (X_(Q)), andvice versa

Following de-multiplexing, both the in-phase and quadrature-phasesymbols are decoded in respective decision device blocks 136 and 138 togenerate symbol decisions I with a ‘I’ and ‘Q’, respectively. Theoriginal sample value I and Q is arithmetically combined with thedecisions ‘I’ and ‘Q’ in respective adders 140 and 142 in order togenerate an error term E_(I) and E_(Q), respectively for the I rail andthe Q rail.

Phase error terms are generated, one for each rail, as P_(I) and P_(Q),respectively by taking the product of a particular rail's error term andmultiplying it by the corresponding rail's mid-symbol point. Forexample, the phase error term for the I rail, P_(I) is equal to thequantity (I−‘I’)*X_(Q).

Alternatively, the phase error term P_(I) might also be represented as(I−‘I’)*sign(X_(Q)). Similarly, the phase error term for the Q rail canbe expressed as (‘Q’−Q)*X_(I), or (‘Q’−Q)*sign(X_(I)) In either case,the I, ‘I’ and X_(Q) in each I rail phase error term computation havethe same time index, as do the Q, ‘Q’ and X_(i) for each Q rail phaseerror term computation. Thus, there should be a corresponding pair ofP_(I) and P_(Q) phase error terms per symbol which exhibit an offsetequal to the offset between the initial I and Q signals.

In accordance with the exemplary embodiment of FIG. 10, each pair ofP_(I) and P_(Q) phase error signals are further multiplexed, inmultiplexer 148, in order to generate two consecutive phase error terms,for each symbol, which are provided in turn to a loop filter 150. Theloop filter 150 develops control voltage for the loop's VCO 152 whichprovides a timing reference signal for an input phase splitter andderotater 154 which functions as a phase correction block.

The carrier phase loop of FIG. 10, will be understood to include a phasedetector capable of extracting the phase and/or frequency differencebetween the transmitted carrier and received carrier in order toaccurately demodulate received signals.

FIG. 11 is a simplified, block level diagram of a carrier phasedetection and correction loop as it might be implemented if the receiverwere operating in VSB mode. In FIG. 11, the input signal is received bya phase splitter and derotater 154 which provides a single sidebandsignal (I) to the VSB phase detector 156, along with a counterpartsignal X_(Q), where the single sideband signal I and its counterpartX_(Q) form a Hilbert transform pair.

As was the case in connection with the QAM phase detector 130 of FIG.10, the I signal is decoded, i.e., quantized to a valid symbol, in adecision device 158 to generate a valid symbol ‘I’. The ‘I’ and thevalid symbol ‘I’ are negatively combined in a summing circuit 160 inorder to define an error term E_(I). A multiplier 162 combines the errorterm E_(I) with the sideband signal counterpart X_(Q) in order to definea phase error term P_(I) which can be expressed as (I−‘I’)*X_(Q) oralternatively, (I−‘I’)*sign(X_(Q)). The phase error term P_(I) isprovided to a loop filter 164 which develops a control voltage for areference signal synthesizer such as a voltage controlled oscillator(VCO) 166. The synthesized reference is provided, in turn, to the phasesplitter and rotater 154 in order to provide a phase correction toincoming single sideband VSB signals.

Turning now to FIG. 12, there is depicted a simplified, semi-schematicblock diagram of an alternative embodiment of a VSB-type carrier phasedetection and correction system in accordance with the presentinvention. As was the case with the carrier phase detection andcorrection system of FIG. 11, the system of FIG. 12 receives a singlesideband (VSB) signal through a phase splitter and derotater 170 andprovides a I signal to a summing junction 172 where it is combined withthe output of a decision feedback equalizer (DFE) 174. It should benoted herein that the summing junction 172 and DFE 174 correspond to thesumming junction 117 and DFE 118 of the exemplary embodiment illustratedin FIG. 9.

I rail signals are directed to a decision device 176, such as a slicer,where the incoming I signals are quantized to a valid constellationpoint ‘I quantized symbols, i.e., decisions, are fed back into the DFE174 and further provided to a second summing junction 178 where they arenegatively combined with the input signal I in order to define an errorterm E_(I) representative of the displacement of the input signal I fromits ideal (quantized valid) value. The error signal E_(I) may thus beviewed as representing a rotational or phase error of the input signal Ifrom its ideal quantized value ‘I.

In order to determine the direction of phase rotation, i.e., todetermine a phase lead or phase lag, the error E_(I) is multiplicativelycombined with a midpoint signal X_(Q) which is the Hilbert transform ofthe input signal I. X_(Q) is developed through a Hilbert transformcircuit 180. X_(Q) might be combined directly with the error term E_(I)in a multiplier 182 or alternatively, might be evaluated to determineits sign in an optional sign determination circuit 184. Thus, the outputof the exemplary carrier phase detector of FIG. 12 is a generated phaseerror term is P_(I) which might be expressed as either (I−‘I’)*X_(Q) or(I−‘I’)*sign (X_(Q)). The phase error term P_(I) is directed to a loopfilter 186 which develops a control signal (a control voltage) thatcontrols the operational parameters of a reference signal synthesizersuch as a VCO. The synthesized reference signal is provided to the phasesplitter and derotater 170 which, in turn, “derotates” incoming signalsin order to properly recover and track carrier phase.

It should be understood that the exemplary embodiment of FIG. 12, wherethe Hilbert transform X_(Q) of an incoming I signal is generatedinternally, is suitably implemented in the case where the phase detectoris provided in combination with a baseband decision feedback equalizer(DFE) or when the derotater is implemented such that only an in-phasecomponent of a received signal is produced. In the exemplary embodimentof FIG. 11, a Hilbert transform circuit would not be required since theHilbert transform X_(Q) of the incoming signal I is directly available.

Returning to FIG. 12, it should be understood that the Hilbert transformcircuit 180 introduces some measure of delay in the signal path betweenthe input to the decision device 176 in the input to the multiplier 182for combination with an error term E_(I). An arbitrary delay introducedin any one leg of the signal paths would thus contribute a delay term(an additional error term) to the output phase error term P_(I) whichwould have the effect of either over or under compensating any phaselead or phase lag exhibited by the incoming signal. Accordingly, anadditional delay stage 190 is introduced between the output of thesumming junction 178 and the multiplier 182 in order to match the delayintroduced by the Hilbert transform circuit 180. Since the phasedetection and correction circuit of FIG. 12 is immanently suitable forimplementation in integrated circuit technology, the delays caused bythe Hilbert transform circuit 180 can be calculated to a reasonabledegree of accuracy. Once delay has been characterized, a suitablematching delay can be devised by constructing the delay stage 190 withsimilar integrated circuit components having similar responsecharacteristics and parasitic resistances and capacitances to theHilbert transform circuit 180.

It should be understood that in carrier recovery systems based on thepilot, both QAM and VSB constellations are able to be accurately decodedso long as the phase of the pilot accurately represents the averagephase of the signals. However, as is well understood by those havingskill in the art, typical communication channels exhibit a non-linearphase response causing the pilot to often be attenuated. The channelphase response at the pilot location is quite often different from thechannel phase response elsewhere along the spectrum, thus causing aconstellation to be effectively rotated when evaluated in connectionwith pilot phase. Since a systems' equalizer is expecting true basebandfrom the pilot, the system response might be accurately characterizedwith respect to pilot frequency but not necessarily accurately withrespect to pilot phase, i.e., the system exhibits pre-equalizerrotation. Since the Equalizer is expecting to receive a signal thatmight be characterized as A(t)e^(jwt+φ), where the first portion of theexponential term represents frequency and the second portion of theexponential term represents phase.

In order to minimize pre-equalizer rotation, the dual mode QAM/VSBreceiver according to the invention incorporates a 1-tap LMS derotaterin OQAM mode in order to perform a pre-equalizer phase correction.Turning now to FIG. 13 and with reference to the exemplary embodiment ofa dual mode QAM/VSB receiver of FIG. 9, the 1-tap LMS derotater issuitably disposed in the signal path before the equalizer 24 (alsoidentified with the same reference numeral in the exemplary embodimentof FIG. 9). Complex signals I and Q are evaluated in a decision device200 that is implemented in the exemplary embodiment of FIG. 13 as aslicer. Input complex signals I and Q are negatively combined withquantized OQPSK symbol values in a summing junction 202 in order todefine a complex error term E. Complex error E is processed by a leastmeans squares (LMS) function block 204 to develop a rotational alpha αhaving the conventional representation α_(n+1)=α_(n)−μ·X*·E, where μrepresents the step change.

α is applied to a complex multiplier 206 where it is used to provide anyneeded pre-equalizer rotation correction before the complex signals Iand Q are directed to the equalizer circuit 24.

As discussed above in connection with FIG. 9, the dual mode QAM/VSBreceiver according to the invention incorporates a decision feedbackequalizer (24 of FIGS. 1 and 9) which is suitably constructed of afeedforward filter section (or FFE) and a decision feedback filtersection (or DFE) as is illustrated in the simplified, semi-schematicexemplary embodiment of FIG. 14. In particular, an input signal,represented as x_FFE_in is directed to a feedforward filter element 200.After filtering, the signal is summed with the negative of the output ofa decision feedback filter element 202 in a complex summing junction204. Summed signals are directed to the input of a decision device 206,such as a multi-level slicer, which provides a signal decision, denotedx_dec, at one output and an error term, denoted “error”, representing avector difference between a valid, quantized constellation point anactual received value. Decisions developed by the slicer 206 are furtherdirected to the input of a decision feedback filter element (DFE) as aparallel signal denoted a DFE word, and identified as dfe_w. Thus, as iswell understood by those having skill in the art, the received signalx_ffe_in is only used by the feedforward filter element 200, while anestimated decision signal x_dec is used by the decision feedback filterelement 202. At the summing junction 204, an FFF filter version of thetail portion of the channel impulse response is canceled by the DFFweighted estimated signals. Any noise enhancement introduced by a DFE isonly caused by equalization of the remaining smaller portion of thechannel impulse response. As is also well understood, a feedforwardfilter (FFF) compensates for channel distortion with linear equalizationand can be implemented at multiples of the baud rate. Decision feedbackfilters (DFF) cancel the tail portion of the channel impulse responseusing recovered data symbols. As such, a DFF can be implemented only atthe baud rate.

No matter how implemented, the decision feedback filter element 202 is ahighly complex system which performs a significant number of arithmeticcalculations, at extremely high clocking speeds. The number ofcalculations performed necessarily depends upon both the length of thefilter, i.e., the number of coefficients (or taps) that contribute tothe final output signal, as well as the width of the filter, representedby the filter's wordlength or the number of bits required forrepresentation of the symbols at the input of the DFE. Reducing thewordlength of the decision feedback filter 202 significantly reduces thecomplexity of the decision feedback filter block.

FIG. 15 is a simplified block level diagram illustrating an exemplary8-tap decision feedback filter, suitably implemented as a sequentialstring of delay stages, with the output of each delay stage, as well assymbols at the DFF input, each being multiplied by a correspondingcoefficient, denoted d(0) . . . d(7). Each of the coefficient multipliedsignals are summed together at summing junctions in order to define afilter output y(n). It should be understood that d(0) . . . d(7) denotes8 multiplication operations, each of which require significantinvestments in processing hardware, and each of which are performed inparallel fashion with an index equal to the decision feedback filterword length dfe_w. The hardware complexity of these multiplicationoperations are linearly reduced when the word length dfe_w is reduced.

QAM modulated signals include an in-phase component and aquadrature-phase component, denoted I and Q respectively, which requiresthe use of a complex decision feedback equalizer such as depicted insemi-schematic block diagram form in FIG. 16. Briefly, the exemplarycomplex decision feedback equalizer of FIG. 16 includes real andimaginary filters for each of the in-phase and quadrature-phase inputcomponents. For example, an in-phase signal Iin is processed by a realfilter 208 whose output is summed with the output of an imaginary filter212 which receives a quadrature-phase input. Likewise, thequadrature-phase input is processed by a real filter 214 whose output issummed with the output of an imaginary filter 210 which, in turnreceives an in-phase signal Iin as an input. Thus, it should beunderstood that the exemplary complex decision feedback equalizer ofFIG. 16 is nothing more than a graphical, block diagram representationof the mathematical function defining a complex filter.

In the case where the exemplary complex decision feedback equalizer ofFIG. 16 is operating on a 256 QAM signal, 8-bits are required toadequately define the representation of each symbol at the input to theexemplary complex dfe. This is because 256 QAM symbols requiresLOG₂(256)=8-bits for symbol representation. The 8-bit representation ofeach symbol can be divided into two subsets, with 4-bits chosen torepresent real symbols (denoted as in-phase or I symbols) and 4-bitschosen to represent imaginary symbols (denoted as quadrature-phase or Qsymbols). When represented in this manner, a 256 QAM constellation mightappear as depicted in the graphical representation of FIG. 17, andsuitably includes the 2⁴=16×16=256 complex symbols, symmetricallyarranged about the I and Q axis.

In order to maintain the symmetry about the zeros of the I and Q axis,as well as for ease of numerical processing, the binary two's complimentnumbering system is used for implementing the signal processingfunctions in the exemplary dual mode QAM/VSB receiver. Utilizing two'scompliment as the numbering system, results in a −½ bit offset inrepresentation of each of the QAM symbols. As can be determined from theexemplary 256 QAM constellation represented in FIG. 17, quantized symbolpoints (desired symbols) range from −{fraction (15/16)} to {fraction(15/16)} on both the I and Q axes with a ⅛ offset, or separation,between symbol points. Thus, an input signal x(n) would take on valuesof {−{fraction (15/16)}, −{fraction (13/16)}, −{fraction (11/16)},−{fraction (9/16)}, −{fraction (7/16)}, −{fraction (5/16)}, −{fraction(3/16)}, −{fraction (1/16)}, {fraction (1/16)}, {fraction (3/16)},{fraction (5/16)}, {fraction (7/16)}, {fraction (9/16)}, {fraction(11/16)}, {fraction (13/16)}, and {fraction (15/16)}}.

However, a 4-bit representation of each of the 256 QAM symbol points inthe two's compliment numbering system can be expressed as{100_(b),1001_(b),1010_(b),1011_(b),1100_(b),1101_(b),1110_(b),1111_(b),0000_(b),0001_(b),0010_(b),0011_(b),0100_(b),0101_(b),0110_(b),0111_(b)} which, whenexpressed in common numerical form represents an input signal, denotedby z(n), which takes on the discrete values {{fraction (16/16)},−{fraction (14/16)}, −{fraction (12/16)}, {fraction (10/16)}, −{fraction(8/16)}, −{fraction (6/16)}, −{fraction (4/16)}, −{fraction (2/16)}, 0,{fraction (2/16)}, {fraction (4/16)}, {fraction (6/16)}, {fraction(8/16)}, {fraction (10/16)}, {fraction (12/16)} and {fraction (14/16)}}.Thus, it will be understood that the effective input signal z(n), whenprocessed, would give symbol quantization results that are incorrect bya fixed offset, equal to −{fraction (1/16)}, and which is denoted hereinby a. Since −{fraction (1/16)} may be represented by the binary value00001 in two's compliment, the −{fraction (1/16)} fixed offset may becorrected by adding a=00001 to z(n) as a correction factor.

While effective to some degree, adding a correction factor in thismanner raises the number of bits required to represent each I and Qsymbol from 4-bits to 5-bits for each discrete symbol point. Thus, aninput signal x(n) that accurately represents discrete symbol pointswould be represented by {10001, 10011, 10101, 10111, . . . 01001, 01011,01101, and 01111} which correctly represents the desired discrete symbolvalues from −{fraction (15/16)} to {fraction (15/16)}. However,increasing the wordlength required to accurately represent a symbol from4-bits to 5-bits linearly increases the complexity of the multipliersused to implement the decision feedback filter portion of the system'sDFE.

In accordance with the invention, a decision feedback equalizer (DFE)constructed in accordance with the simplified, block diagram of FIG. 18,includes a decision feedback filter 220 that accommodates a two'scompliment representation of discrete symbol points in a manner thatminimally effects the increase in hardware complexity caused by thetwo's compliment numbering system. In the exemplary embodiment of theDFE of FIG. 18, an input signal x_ffe_in is received by a feedforwardfilter 222, having a coefficient set represented by f(n). Thefeedforward filter's output is directed to a decision device 224, suchas a slicer, configured to output a tentative decision, represented byx_dec and an error term. Tentative decisions are directed to the inputof a decision feedback filter 220 in the form of an input signal,denoted x(n), which may be alternatively described as a DFE wordrepresented as dfe_w. However, and in accordance with the invention, theDFE word is a 4-bit representation and, thus, does not include the fifthbit fixed offset term denoted by a. Thus, the DFE word is represented inthe exemplary embodiment of FIG. 18 as dfe_w-a.

Returning momentarily to the exemplary decision feedback filter of FIG.15, it will be realized that such a filter, with input x(n), outputy(n), and coefficients d(n) can be characterized by the convolutionalequation y(n)=Σ_(k)d(k)x(n−k). However, if the input signal to theexemplary filter of FIG. 15 is viewed as including an input stimulusportion and a fixed offset portion, the input signal could be expressedas x(n)=z(n)+a. Given this particular mathematical relationship, andsubstituting terms in the filter response characteristic, the filteroutput might be represented as y(n)=Σ_(k)d(k)z(n−k)+Σ_(k)d(k)a, where arepresents the fixed offset term and z(n)represents the input stimulus.

From the above, it will be evident that the filter's output might beexpressed as the sum of two independent terms, a first term in which theinput stimulus z is convolved with the coefficient set d and a secondterm in which the coefficient set d is convolved with the fixed offsetterm a. When separated, the first portion of the filter's responsecharacteristic retains the original representational word length (4-bitsaccording to the foregoing exemplary description) which is directed tothe input of the decision feedback filter 220 of the exemplaryembodiment of the DFE of FIG. 18. The characteristic filter output y(n)is developed in a summing node 226 which sums the output of the decisionfeedback filter 220 with the output of an offset generation circuit 228which provides an offset correction signal equal to the convolution ofthe decision feedback filter coefficients d with the fixed offset terma. Thus, an offset equal to Σ_(k)d(k)a is added to the output of thedecision feedback filter 220 at the summing junction 226 down-streamfrom the output of the decision feedback filter.

The offset cancellation circuit 228 might be constructed as a simpleregister which receives adaptively defined coefficients d(n) from thedecision feedback filter 220. Coefficient values are multiplied by thefixed offset a and summed for all k to define the offset term added tothe output of the decision feedback filter. Accordingly, intensivenumerical processing, performed by the decision feedback filter 220, isperformed on a DFE wordlength of only 4-bits. Processing required togenerate the offset term is minimal and requires a significantly lowerinvestment in computational hardware than if the offset term wereincorporated in the DFE word as a fifth bit. Computational complexity issignificantly reduced as a consequence.

The decision feedback filter output y(n), which includes the offsetterm, is negatively summed with the output of the feedforward filter 222at a summing circuit 230. The sum of the negative of the decisionfeedback filter output and the feedforward filter output is provided asan input to the slicer 224. It bears mentioning that the error termdeveloped by the slicer 224 is provided as a control input to both thedecision feedback filter 220 and the feedforward filter 222 foradaptively modifying the content of the coefficient registers such thatthe decision feedback filter and feedforward filter converge. Needlessto say, the adaptively reconfigured coefficients d(n) of the decisionfeedback filter 220 are provided to the offset cancellation circuit 228to accurately correlate the offset term to the output of the decisionfeedback filter. The extent of the reduction in computational complexityof the filters in a DFE constructed in accordance with the inventionmight be better understood when it is recognized that the exemplary dualmode QAM/VSB receiver includes a 496-tap decision directed equalizerhaving 64 feedforward taps and 432 feedback taps. A 20% reduction in thedecision feedback filter circuitry (reducing the DFE wordlength from5-bits to 4-bits) more than compensates for the minimal additionalhardware required by the offset cancellation circuit 228.

The foregoing discussion was concerned with reducing the computationalcomplexity of DFE elements that might otherwise have obtained as aresult of carrying forward a fixed offset value of a two's complimentrepresentation of a 256 QAM constellation. In the case of VSB modulatedsignals, the same type of symbol representational offset occurs withregard to a VSB constellation, as well as a DC offset term introduced asan artifact of the above-described pilot. The ATSC standard for VSBtransmission requires utilization of a pilot tone as a carrierreference. When a received spectrum is mixed down to baseband, the pilottone reduces to a DC component at baseband and which must consequentlybe subtracted from an equalizer signal prior to its introduction to theslicer and subsequently added back to the decision signal defined at theslicer output.

FIG. 19 is a simplified block level diagram of a DFE similar to theexemplary embodiment of FIG. 18, but further including a pilot tonegeneration circuit 232 which functions to develop a DC component equalto the pilot's DC component baseband. The DC component developed by thepilot tone generation circuit 232 is negatively summed with the filteredDFE input in a summing circuit 234 prior to the signals being directedto the slicer 224. The DC component developed by the pilot tonegeneration circuit 232 is further added to the tentative decision x_decdeveloped by the slicer, in summing circuit 236, prior to the decisionsbeing provided to the input of the decision feedback filter 220 and alsoprior to its being output from the DFE for decoding and errorcorrection. The DC component, thus added to the decision value has theeffect of increasing the DFE wordlength at the input to the decisionfeedback filter 220. In the case of 8 VSB the discrete symbol valuesmight be represented as {−⅞, −⅝, −⅜, −⅛, ⅛, ⅜, ⅝, and ⅞} with the valueof the pilot tone generally recognized as {fraction (5/32)}. In ordinarybinary terms, the DFE wordlength that would be necessary to represent 8VSB symbols is 3-bits. However, utilizing the two's compliment numberingsystem results in a −⅛ offset, in a manner similar to the QAM casedescribed above, which requires an additional bit for itsrepresentation, resulting in 4-bits being required to accuratelyrepresent each of the 8 VSB constellation points. When the pilot tonevalue is factored into the foregoing, it should be understood that thepilot tone further increases the DFE wordlength by adding {fraction(5/32)}, or 000101_(b) in two's compliment representation, to eachsymbol value, resulting in a 2-bit increase to the wordlength (thusincreasing the wordlength from 4-bits to 6-bits). Thus, the DC pilottone component of {fraction (5/32)}, in combination with the fixed −⅛symbol offset, necessitates an approximately 50% increase in thecomputational complexity of a decision feedback filter of a DFEoperating on 8 VSB modulated signals.

Turning now to FIG. 20, an additional exemplary embodiment of a decisionfeedback equalizer, configured for VSB modulated signals, andconstructed to reduce DFE wordlength from the nominal 3-bit case back toan original 3-bit representation, is illustrated in simplified,semi-schematic block diagram form. In particular, the pilot tonegeneration circuit 232 develops a DC component equal to the pilot toneDC component at baseband, and provides the component value to a summingcircuit 240, where it is subtracted from the filtered DFE signal priorto its introduction of the slicer 224. However, and in accordance withthe invention, the pilot tone generation circuit 232 is decoupled fromthe decision output of the decision slicer 224 and instead provides theDC component corresponding to the pilot tone to an offset correctioncircuit 242 which functions to compensate the output of the decisionfeedback filter 220 with an offset term a equal to the value of thepilot tone minus the ⅛ offset introduced by the two's complimentnumbering system.

In general terms, the same mathematical analysis may be performed on theDFE exemplified in FIG. 20 as was performed on the DFE exemplified inFIG. 18. Specifically, the tentative decision x_dec is provided to thedecision feedback filter 220 as a 3-bit DFE word, dfe_w-a, that does notinclude the 2-bits representing the −⅛ computational offset and the{fraction (5/32)} pilot tone value. The pilot tone and computationaloffset values are convolved with the filter's coefficient values inorder to define an offset term which is summed with the filter output ina summing circuit 244 to define a decision feedback filter output y(n).The DFF output y(n) is subtracted from the output of the feedforwardfilter 222 in the summing circuit 230 prior to its introduction to theslicer 224.

Not only does the offset correction circuit 242 function tosignificantly reduce the complexity of the decision feedback filter 220in the VSB case, but it also allows the pilots on generation circuit 232to be decoupled from the slicer's output. In the conventional DFEembodiment of FIG. 19, the pilot tone value is added back to the sliceroutput prior to the tentative decision's being provided to the decisionfeedback filter because the pilot tone value is subtracted from theslicer's input signal, after the decision feedback filter output hasbeen combined with the feedforward filtered input signal. The DCcomponent removed from the signal after the decision feedback filter 220must be replaced in order that the filter remain converged.

In the exemplary embodiment of FIG. 20, pilot tone DC compensation, aswell as computational offset computation, occurs in a loop disposedinside the feedback loop of the decision feedback filter, as well asoccurring prior to the slicer 224. Thus, pilot tone DC compensationoccurs twice in the signal path between the decision feedback filteroutput and the input to the slicer; a first compensation associated withthe offset correction circuit 242, where a compensation term related tothe pilot term is added to the output of the decision feedback filter,and a second compensation occurring just prior to the input of theslicer where the pilot tone DC component is removed from the inputsignal. Completely removing the pilot tone DC component from the signalwithin the DFE is further advantageous in that there is not DC offsetpresent in signals provided to the decoder and forward error correction(FEC) circuitry following the DFE. Additional savings in the complexityof FEC and decoder circuitry can be realized by obviating therequirement that a signal from the DFE be processed with DC offsetcomponents.

Trellis coded modulation is employed in modern digital communicationsystems to improve a system's bit error rate in high noise situations.Trellis coded modulation (TCM) achieves a performance gain by increasingthe size of a constellation within the modulation scheme, therebyincreasing the “distance” between possible transmitted sequences. Aparticular example of a TCM communication system might include the U.S.digital terrestrial broadcasting standard, which employs a trellis coded8 VSB modulation scheme. The particular code used has an asymptoticcoding gain of 3.31 db over uncoded 4 VSB.

FIG. 21 is a simplified, semi-schematic block diagram of an exemplaryencoder which might be provided in a typical terrestrial broadcasttransmitter, and which might be represented in simplified form as aconvolutional encoder 300 in combination with a signal mapper 302. A2-bit input signal, Y₁ and Y₂ are input to the convolutional encoder 300with the least significant bit, Y₁, also directed, in parallel fashion,through a convolutional encoder, implemented as a linear feedback shiftregister, in order to generate a redundancy bit which is a necessarycondition for the provision of coding gain of the code.

As described above, the convolutional encoder 300 includes a linearfeedback shift register, constructed of two delay elements 304 and 306(conventionally denoted by Z⁻¹) separated by a summing circuit 307,which function to combine the least significant bit Y₁ of the input wordwith the output of the delay elements 304 and 306. The time sequenceformed by the LSB bit stream is convolved with the coefficients of thelinear feedback shift register in order to produce the time sequence ofthe redundancy bit. Thus, the convolutional encoder might be viewed as astate machine.

The signal mapper 302 maps the resulting 3 bits, Z₂Z₁, and Z₀ into aparticular constellation level. Since there are 3-bits coming into thesymbol mapper 302, a maximum of 8 levels might be represented bycombinations of the 3-bits. As will be understood from the block diagramof FIG. 21, the 8 possible levels might be represented as −7, −5, −3,−1, 1, 3, 5 and 7 .

However, since coding increases signal modulation from 4 levels to 8levels, decision directed loops, such as decision directed adaptiveequalization, decision directed carrier and/or timing recovery loops,and the like, are forced to function with respect to an increasedconstellation size of 8 levels.

Turning now to FIG. 22, an exemplary decision directed carrier andtiming recovery loop is shown in simplified, semi-schematic blockdiagram form, and includes a symbol-by-symbol slicer 310 as a decisiondevice, operating in combination with a DFE 312 to generate tentativedecisions suitable for use by a carrier loop 314 and timing loop 316.However, at signal-to-noise ratios (SNR) near system threshold, theloops will fail due to the combination of higher noise and largerconstellation size. As a result, the system will not be able to achieveadequate lock, and the expected coding gain from TCM would not berealized. In particular, a symbol-by-symbol slicer does not employsequence estimation in generating symbolic decisions. Rather, itoperates only upon the “current” symbol, ignoring any past decisions.

On the other hand, were the DFE input to be taken from a best survivorpath in a trellis decoder's trace back memory, the system would be ableto exploit the correlations between a “current” symbol and pastdecisions, by maximum likelihood sequence estimation, for example. TheDFE input would thus exhibit a lower error rate and, with a higherpercentage of correct decisions, the DFE's ability to operate in low SNRenvironments is enhanced.

Turning now to FIG. 23, there is shown in simplified, semi-schematicblock diagram form, a generalized decision feedback equalizer circuitthat includes a TCM demodulation circuit, also termed a Viterbi decoder,which provides the input to a decision feedback equalizer 322. Thesystem includes a carrier loop 324 that drives a derotater 326 disposedbetween the Viterbi 320 and a feedforward equalizer 328.

In addition to the carrier loop 324, the system also includes a symboltiming loop 330, coupled to provide a symbol timing reference to avariable interpolating filter 332. Although the symbol timing loop 330is depicted in the exemplary embodiment of FIG. 23, the symbol timingloop 330 need not be decision directed, in the context of the presentinvention, but might alternatively be configured to operate upon anenhanced pilot signal in a manner described in connection with FIGS. 4and 9.

In accordance with the invention, the input and output of the Viterbi320 is directed to a summing junction 334 which combines an input signaland a tentative decision from the Viterbi in order to generate an errorterm. The error term, in turn, is used to drive the coefficient tapupdate of the FFE 328, as well as the coefficient tap update of the DFE322. Providing a lower probability of error in the tap update signalsignificantly improves the performance and reliability of the FFE 328.

As will be described in greater detail below, TCM decoders exhibit atradeoff between system delay and the reliability of symbolic decisions.In general, making use of decisions farther back in the history of a TCMdemodulator tends to increase the reliability of the decision, with mostreliable decision being the final decision. However, each stage in theprocess involves a certain amount of delay and it is sometimes desirableto choose decisions from some intermediate point of the tracebackhistory. The earlier the chosen decision, the less the consequent delay.Accordingly, variable delay circuits 336 a, 336 b and 336 c are providedbetween the input of the Viterbi 320 and the summing junction 334, thecarrier loop 324 and the timing loop 330. The variable delay circuits336 a, b and c function to match the delay of the chosen symbol outputfrom the Viterbi such that the summing junction 334, carrier loop 324and timing loop 330 operate on signals having the same time stamp.

Turning now to FIG. 24, a TCM decoder, or Viterbi decoder, is depictedin semi-schematic, block diagram form at 320. A Viterbi suitablyincludes a decision device 340 coupled to receive an input signal froman FFE 328 that has been summed with the output of a DFE 322 in asumming junction 342. A Viterbi decoder processes information signalsiteratively, tracing through a trellis diagram corresponding to the oneused by the encoder, in an attempt to emulate the encoder's behavior. Atany particular time frame, the decoder is not instantaneously aware ofwhich node (or state) the encoder has reached. Thus, it does not try todecode the node at that particular time frame. Instead, given thereceived sequence of signal samples, the decoder calculates the mostlikely path to every node and determines the distance between each ofsuch paths and the received sequence in order to determine a quantitycalled a path metric.

Further, the Viterbi 320, in accordance with the invention, makes anassumption that the surviving paths at the Nth time frame pass through acommon first branch and outputs a decision for time frame 0 on the basisof that assumption. If this decision is incorrect, the Viterbi 320 willnecessarily output a few additional incorrect decisions based on theinitial perturbation, but will soon recover due to the nature of theparticular relationship between the code and the characteristics of thetransmission channel. It should be noted, further, that this potentialerror introduction source is relatively trivial in actual practice,since the assumption made by the Viterbi that all surviving paths attime frame n pass through a common first branch at time frame 0, as acorrect one to a very high statistical probability.

In FIG. 24, the exemplary trellis decoder (or Viterbi) 320 furtherincludes a path metrics module 344 and a path memory module 346 inaddition to the decision device 340. A path metric, as the term is usedherein, is well known and refers to a plurality of elemental pathsbetween neighboring trellis nodes, which form, by extension, a path. TheViterbi selects the best path for each incoming signal and updates apath memory stored in the path memory module 346 and the path metricsstored in the path metrics module 344. It will, thus, be understood thatthe path (or trace) memory module 346 includes a historical record of aparticular number of past decisions, with the number of past decisionsrepresented by a depth parameter N.

Any one of a number of historical decisions may be taken from the pathmemory 346 and provided both to the DFE 322 and an error term generatingsumming junction 334 by selecting the appropriate historical signalthrough a multiplex circuit 348.

Turning now to FIG. 25, the TCM demodulator (or Viterbi) 320 might beconsidered as including 4 traceback registers, with each tracebackregister specific to a particular one of the 4 states making up the 8VSB signal. A MUX 348 selects one of the 4 traceback registers,corresponding to the one containing the most likely symbol, inaccordance with a select signal defined by the path metrics module (344of FIG. 24). The particular symbolic decision chosen by the MUX 348, isoutput from the TCM demodulator and provided to the DFE 322 where it iscombined with a set of N non-causal coefficients, where N represents thelength N of each of the traceback registers. Further, the outputsymbolic decision from the TCM demodulator 320 is processed by a set ofM+1 causal coefficients in the DFE 322, where M represents thedifference between the total number of coefficient taps and the lengthof the traceback register (the number of non-causal taps).

Further, the output of the TCM demodulator 320 is provided to a summingjunction 334 where its value is combined with the TCM demodulator inputin order to define an error term based upon the difference between aninput signal sample and an output symbolic decision. This error term isthen provided to both the DFE 322 and an FFE 328 where it is used toupdate the tap coefficients.

As was mentioned earlier, symbolic decisions may be taken from each ofthe traceback memories at any one of the intermediate steps in theprocess. Depending upon the sequential position of the actual symbolicdecision tap, a certain delay can be determined and that amount of delayis accommodated in a delay circuit 350 disposed between the input of theTCM demodulator 320 and the summing junction 334 in order that the timestamp of the input signal and the time stamp of the symbolic decision tobe summed are equal. This delay is variable and programmable in thatcircuit simulations may be run in order to determine thedelay/performance tradeoff characteristics. Either performance or delay(or a mixture of both) might be set as a decision metric and the systemoptimized for either maximum performance, minimum delay, or an adequatevalue of both. It is indicated in the exemplary embodiment of FIG. 23,the symbolic decisions, and consequent delay, need not necessarily bethe same for defining the error term, providing an input to the carrierloop or the timing loop. Indeed, because of the different bandwidthconstraints and acquisition characteristics of a carrier loop and atiming loop, it should be understood that the carrier loop needs toacquire at a much faster rate than the timing loop, allowing the timingloop to use a more “downstream” survivor path in the trellis decoder'spath memory module and not be too concerned with its attendant delay.

In the case of a carrier loop, using decisions farther back in thehistory of the TCM decoder would tend to increase the reliability ofdecisions. However, increasing delay in the carrier loop correspondinglyreduces the loop's tracking ability. Thus, the variable delay feature ofthe invention enhances overall system performance of a multi-loopdecision directed system, as well as providing improved equalizationcharacteristics.

The foregoing discussion discloses and describes merely exemplarymethods and embodiments of a dual mode QAM/VSB receiver in accordancewith the present invention. As will be understood by those familiar withthe art, the various features and functions of the invention may beembodied in a variety of other specific forms without departing from thespirit or essential characteristics thereof. Accordingly, the disclosureof the present invention is intended to be illustrative of, but notlimiting to, the scope of the invention, which is set forth in thefollowing claims.

What is claimed is:
 1. A method for operating a receiver having aparticular sampling frequency related to a symbol rate to recovercarrier and timing information from a received spectrum including apilot signal, wherein the pilot signal is provided at a predeterminedfrequency, the method comprising: centering the received spectrum at aknown position relative to baseband; tracking the pilot signal with aphase-lock-loop; evaluating the frequency of the pilot signal withrespect to the particular sampling frequency; and adjusting thecentering of the received spectrum until the evaluated frequency of thepilot is in integral relationship with the particular samplingfrequency.
 2. The method according to claim 1, further comprising:filtering the received spectrum in a low pass filter having a cut-offfrequency related to the sampling frequency; processing the filteredsignal in a high pass filter having a cut-off frequency related to thesampling frequency; and wherein the low pass and high pass filtersdefine an equivalent bandpass filter, the equivalent bandpass filterdefining upper and lower sideband regions each centered at an expectedposition of the pilot signal.
 3. The method according to claim 2,wherein the received spectrum exhibits a raised cosine responsecharacteristic, the upper and lower sideband regions of the equivalentfilter including transition regions of the spectra defined by the highpass and low pass filters.
 4. The method according to claim 3, whereinthe receiver's sampling frequency is determined such that the pilotfrequency is equal to about one-fourth the sampling frequency.
 5. Themethod according to claim 4, wherein the high pass filter is a Nyquistprefilter having a lower cut-off frequency at about one-fourth thesampling frequency, the high pass filter centering the spectrum at afrequency about one half the sampling frequency.
 6. The method accordingto claim 5, wherein the low pass filter is a square root Nyquist filterhaving an upper cutoff frequency at about one-fourth the samplingfrequency, the square root Nyquist filter centering the spectrum atabout baseband.
 7. The method according to claim 6, the desired positionof the pilot signal being substantially centered in each transitionregion, thereby causing the desired position of the pilot signal to becentered in each of the upper and lower sideband regions of theequivalent filter at a frequency substantially equal to one-fourth thesampling frequency, thereby augmenting a signal occurring at one-fourththe sampling frequency.
 8. The method according to claim 7, furthercomprising: receiving a sideband region from the equivalent filter in atracking loop; sweeping the sideband region to identify an augmentedfrequency component; comparing the augmented frequency component to theexpected frequency of the pilot signal; and shifting the samplingfrequency in a direction and an amount such that the augmented frequencycomponent coincides with the expected frequency of the pilot signal. 9.The method according to claim 8, further comprising the step of usingthe shifted sampling frequency to define a symbol timing referencesignal.
 10. In an integrated circuit communication device, a method forsynchronizing a receiver's timebase to a remote transmitter'scomprising: receiving an input spectrum including a pilot signal,wherein the pilot signal is provided at a predetermined frequency;providing a sampling frequency having a fixed relationship with thepilot frequency; evaluating the position of the pilot signal in thefrequency domain with respect to the sampling frequency; and adjustingthe sampling frequency when the pilot signal is at a location in thespectrum different from its expected frequency location.